Direct conversion radio receiving system using digital signal processing for channel filtering and down conversion to base band

ABSTRACT

A radio receiving system includes an orthogonally-modulated waveform detection/channel filter section  4,  an I signal root Nyquist filter  20,  a Q signal root Nyquist filter  21,  and a signal detection/demodulation section  25.  The orthogonally-modulated waveform detection/channel filter section  4  further includes a first filter (a band pass filter)  2  which permits passage of only a signal at a frequency band assigned to a communications system from which the radio receiving system receives a signal; a sample-and-hold circuit  5;  a Hilbert transformer  6;  a first channel filter  7  through N-th channel filter  9;  and a clock signal shaping/controlling section  15.  The band pass filter  2  is provided with the characteristics which cancel the aperture effect due to a sampling operation, thus compensating for the aperture effect due to a sampling operation.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a radio receiving system based on anorthogonal modulation communication method, and more particularly, to aradio receiving system which compensates for an aperture effect due to asampling operation by setting the frequency characteristics of a bandpass filter—which permits passage of only a signal at a frequency bandassigned to a communications system from which the radio receivingsystem receives a signal—are set so as to prevent the aperture effect.

2. Description of the Related Art

A receiver—which is based on a direct conversion receiving method andwhich has a simplified radio section—is realized through use of achannel filter which samples, or subjects to an analog-to-digitalconversion operation, an input signal while the input signal stillremains in a high frequency state before converted into a basebandsignal and which subjects the quantized signal to a stable digitalsignal processing operation having a high degree of accuracy. However,the channel filter suffers from the following four problems.

First, as a result of the sampling operation, the sampling frequencyrenders the frequency characteristics of the overall radio receivingsystem uneven. Consequently, a digitized signal is demodulated at a higherror rate.

Second, in order to highly accurately perform a sampling operation,previous and subsequent stages of the sampling circuit must have highspeed characteristics required to ensure over a considerably widefrequency range the speed performance of the sampling circuit for thepurpose of preventing the aperture effect. As a result, the samplingcircuit has a bandwidth which is considerably wider than the bandwidthof a received signal. In short, in spite of a band pass filter providedin a previous stage in order to limit the bandwidth of the receivedsignal to a predetermined bandwidth, the circuit provided in thesubsequent stage must have a bandwidth which is significantly wider thanthat of the band pass filter. Thermal noise caused by the circuitprovided in the subsequent stage exceeds the amount of that caused in anexisting radio receiving system, which also accounts for an increase inthe error rate.

Third, under the direct conversion receiving method, there is a need toprovide a base band circuit with a function as a substitute for achannel filter which is conventionally provided in an IF stage of theexisting receiver. To this end, it is also necessary for an HF stagewhose filtering is insufficient to maintain a wide dynamic range and awide bandwidth. Still further, there is a need for a filter whichfilters a signal having such a wide dynamic range and a bandwidth.

Fourth, a sampled signal usually includes d.c. components. Since thesignal becomes vulnerable to d.c. noise, drift, or offsets, the signalincluding such noise accounts for a large error rate in the case of aportable cellular phone based on digital modulation.

FIG. 9 shows an example of an existing direct conversion receiver whichuses a bandwidth-limited sampling method. This circuit diagramcorresponds to a direct IF sampling circuit used in a new produce “125MSPS Monolithic Sampling Amplifier AD9101” described in “Analog DevisesConverter Data Book,” 1^(st) edition, Analog Devises Co., Ltd., July,1997. There are descriptions which state “Adoption of the Nyquist theoryenables elimination of an IF frequency and reconstruction of a base bandsignal. For example, a 40-MHz IF signal is modulated by a signal havinga bandwidth of 10 MHz, and a signal to be detected is detected at asampling rate of 25 MSPS.” A 40-MHz IF signal modulated by a signalhaving a bandwidth of 10 MHz is usually detected at a sampling frequencywhich is twice as high as a frequency of 40 MHz. However, since thesignal is limited to a bandwidth of 10 MHz, the IF signal can bedetected at a sampling frequency of 25 MHz by utilization of the“Shannon's Sampling Theorem” according to which the IF signal can besampled at a frequency twice or more as high as a frequency of 10 MHz.

FIGS. 10A to 10D are views showing variations in spectral componentswhen direct conversion reception is performed through abandwidth-limited sampling operation. FIG. 10A shows a desired waveformand adjacent waveforms in a radio frequency band, as well as thecharacteristics of a band pass filter which covers these waveforms. Inthe drawing, fs designates a sampling frequency set to a frequency whichis twice or more as wide as a communications bandwidth or a thebandwidth of a bandwidth-limited filter.

FIG. 10B shows spectral components of the desired waveform and adjacentwaveforms having frequencies converted into a baseband frequency at asampling frequency. The baseband frequency range fBB is in principle thesame as fBW.

FIG. 10C shows the result of extraction of the desired signal through achannel filtering operation, wherein a quantized signal obtained as aresult of a sampling operation is subjected to a digital signalprocessing operation.

FIG. 10D shows the aperture effect caused by the sampling operation atthis time. In other words, the drawing shows spectral components ofwhat-is-called a sampling function. The spectral components havecharacteristics of {sin(πf/fs)}/(πf/fs), and a null occurs at a samplingfrequency fs. Although the desired waveform in the range less than halfthe sampling frequency does not occur at the null point, the waveform isgiven the frequency characteristics which gradually attenuates awaveform toward higher frequencies.

The present invention relates to a receiver circuit having a built-inchannel filter which is used with a radio receiving system assigned anoffset frequency (disclosed in Japanese Patent Application Laid-open No.Hei-9-266452 “Receiving System,” and Japanese Patent Application No.Hei-9-28271 “Receiving System,” both being filed by the applicant of thepresent patent application) and includes a complex coefficient filter.In the channel filter including a complex coefficient filter on whichthe present invention is based, the center of positive and negativefrequency components to be subjected to a complex operation does notnecessarily occur at zero frequency. For this reason, the aperturecharacteristics of the channel filter which cause the center offrequency components to occur at zero frequency make the operationdistorted, resulting in a considerable decrease in the accuracy of theoperation. Further, even if frequency components are shaped so as tohave complete Nyquist characteristics through use of a subsequent rootNyquist filter, the frequency components cannot have the completeNyquist characteristics.

SUMMARY OF THE INVENTION

The present invention has been conceived to solve the foregoing problemin the existing radio receiving system, and the object of the presentinvention is to solve the problem by providing characteristics ofcompensating for the aperture effect due to a sampling operation to aband pass filter disposed in a receiving input stage of a receivercircuit of a radio receiving system, wherein a channel filter isrealized by quantizing a received signal through sampling and bysubjected the thus-quantized signal to a digitized signal processingoperation.

A first aspect of the invention is directed to a radio receiving systemin which a channel filter is formed by quantizing a received signalthrough sampling and by subjecting the thus-quantized signal to adigitized-signal processing operation, wherein a band pass filter havingcharacteristics of compensating for the aperture effect due to asampling operation is provided in an input receiving stage. Use of theband pass filter having the foregoing characteristics enablescompensation for the aperture effect due to the sampling operation.

According to a second aspect of the invention, the radio receivingsystem is characterized by further comprising: a sample-and-hold circuitfor sampling and holding an output from the band pass filter; and anintegrating circuit having a function of integrating the received signalduring a period of sampling operation of the sample-and-hold circuit. Asa result of the radio receiving system being provided with theintegration effect, the energy of a desired waveform signal can beintegrated. Particularly, even in a case where a weak radio wave isreceived and a desired waveform signal is buried in thermal noise in acircuit, a sampling operation being performed at an ordinary voltageenables power to be produced from an input signal only during a periodover which the aperture effect arises. However, the radio receivingsystem according to claim 2 has the effect of being able to double thepower by integrating the received signal while the period over which thereceived signal is integrated is extended.

According to a third aspect of the invention, the radio receiving systemas defined in the second aspect is characterized by the feature thatintegral action time of the integrating circuit can be changed orselected from a plurality of values. As a result, the integral actiontime of the sample-and-hold circuit is changed with respect to a changein the frequency or bandwidth of the input signal, enabling a desiredintegration effect to be accomplished.

According to a fourth aspect of the invention, the radio receivingsystem as defined in the third aspect is characterized by the featurethat the integral capacity of the integrating circuit is made variable.As a result, the integral action time of the sample-and-hold circuit ischanged with respect to a change in the frequency or bandwidth of theinput signal, enabling a desired integration effect to be accomplished.

According to a fifth aspect of the invention, the radio receiving systemas defined in the second aspect is characterized by the feature anintegrating gate function of the integrating circuit is arranged so asto produce a Nyquist signal waveform. As a result, an efficientsample-and-hold circuit can be realized which provides a superiorsignal-to-noise ratio.

According to a sixth aspect of the invention, the radio receiving systemas defined in either second or third aspect is characterized by thefeature that the time constant of the sample-and-hold circuit is set soas to become longer than the sampling frequency. As a result, thermalnoise or random signals can be sufficiently removed from alower-frequency component. Further, diminution of the sampling frequencyresults in a reduction in the power dissipated by the sample-and-holdcircuit or by peripheral circuits connected thereto.

According to a seventh aspect of the invention, the radio receivingsystem as defined in any one of the first to third aspects ischaracterized by further comprising: sampling means which is made of asample-and-hold circuit and which samples the received signal;difference calculation means for calculating a difference between acurrently-sampled signal received from the sample-and-hold circuit and apreviously-sampled signal; and means for calculating a differencebetween the output from the difference calculation means and an outputfrom the band pass filter and inputs the thus-obtained difference to thesample-and-hold circuit. As a result, the radio receiving system has theeffect of being able to prevent originally-undesired components, such astemperature drift of a sampling circuit or d.c. offset of an inputcircuit, from being mixed into a received signal.

According to an eighth aspect of the invention, the radio receivingsystem as defined in any one of the first to third aspects ischaracterized by further comprising: sampling means which is made of asample-and-hold circuit and which samples the received signal; Hilberttransformation means which produces rectangular components from thesample output from the sample-and-hold circuit; difference calculationmeans for calculating a difference between one of the rectangularcomponents received from the transformation means and apreviously-sampled rectangular component of the same type; and means forcalculating a difference between the output from the differencecalculation means and an output from the band pass filter and inputs thethus-obtained difference to the sample-and-hold circuit. As a result,the radio receiving system is effective in removing a d.c. componentcontained in a sample output produced when the sample-and-hold circuitsamples a received signal together with d.c. components ororiginally-unwanted components mixed in the sample output such astemperature drift of the sample-and-hold circuit or d.c. offset of theinput circuit.

A ninth aspect of the invention is directed to a radio receiving systemwhich receives a signal in a direct conversion receiving mode throughuse of a plurality of cascaded channel filters, each including a complexcoefficient filter, wherein the accuracy of operation of the preliminarychannel filter is improved when compared with that of the subsequentchannel filter. As a result, the radio receiving system is capable ofmore efficiently attenuating an adjacent waveform spaced frequenciesaway from a desired waveform, as well as of supplying to a filtercircuit provided on a subsequent stage a signal of desired waveform fromonly the vicinity of which adjacent waveform signals of strong level areremoved. Accordingly, even if the filter circuit provided on thesubsequent stage is manufactured with less precision: that is, thefilter circuit having the insufficient capability of removing waveformsof great level adjacent to the desired waveform, the radio receivingsystem becomes less apt to suffer from problems.

According to a tenth aspect of the invention, the radio receiving systemas defined in the ninth aspect is characterized by the feature that, inorder to improve the accuracy of operation of the channel filterprovided in the previous stage when compared with that of the channelfilter provided in the subsequent stage, the capacitance of a capacitorwhich is a constituent element of the channel filter provided on thesubsequent stage is set so that the capacitor can be manufactured withhigh dimensional precision, and the capacitance of a capacitor which isa constituent element of the channel filter provided on the previousstage is set so as to become smaller than the total capacitance of thepreviously-described capacitors when they are cascaded. As a result, theprecision of the channel filter is improved through use of capacitorshaving capacitance realized with the highest possible precision.Further, using capacitors having improved dimensional accuracy, acapacitor whose capacitance is smaller than the total capacitance of thecapacitors can be realized.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing the configuration of a receivercircuit of a radio receiving system according to a first embodiment ofthe present invention;

FIG. 2 is a block diagram showing the basic configuration of a channelfilter including a complex coefficient filter common to embodiments ofthe present invention;

FIG. 3 is a schematic representation for explaining the frequencycharacteristics of a single-stage complex coefficient filter common tothe embodiments;

FIGS. 4A to 4D are plots showing the theoretical characteristics valuesof the basic configuration of the channel filter comprising the complexcoefficient filter common to the embodiments;

FIGS. 5A to 5C are schematic representations for explaining phaserotations in respective stages of the complex coefficient filter commonto the embodiments;

FIG. 6 is a plot showing the relationship between the amount of phaserotation occurring in respective stages of the complex coefficientfilter common to the embodiments and a total amount of phase rotation Aoccurring throughout the channel filter;

FIGS. 7A to 7C are plots showing the general frequency characteristicsof two cascaded channel filters, each of which includes the complexcoefficient filter common to the embodiments, wherein when a samplingfrequency of a channel filter provided in a subsequent stage isdiminished to one quarter the original frequency;

FIG. 8 is a block diagram showing a specific configuration of the twocascaded channel filters, each including the complex coefficient filercommon to the embodiments;

FIG. 9 is a block diagram showing the configuration of a radio sectionof an existing radio receiving system which uses a bandwidth-limitedsampling method;

FIGS. 10A to 10D are explanatory views showing a frequency pattern andthe influence of a sampling operation on the frequency pattern in a casewhere a receiving device uses a channel filter including the complexcoefficient filter according to the first embodiment;

FIG. 11 shows specific circuit configurations of orthogonally-modulatedwaveforms separation circuit which are based on Hilbert transformationand which are common to the embodiments;

FIGS. 12A to 12I are timing charts showing timing at which theorthogonally-modulated waveforms separation circuit based on Hilberttransformation and common to the embodiments operate;

FIG. 13 is a circuit diagram showing a specific circuit configuration ofa radio receiving system according to a second embodiment of the presentinvention;

FIGS. 14A to 14C are plots for explaining the operation of the radioreceiving system according to the second embodiment;

FIG. 15 is a view showing the characteristics of a band pass filterwhich compensates for the aperture effect due to a sampling operation;

FIG. 16 is a circuit diagram showing the configuration of a variabledelay device according to a third embodiment of the present invention;

FIG. 17 is a circuit diagram showing a specific circuit configuration ofa radio receiving system according to a fourth embodiment of the presentinvention;

FIG. 18 is a circuit diagram showing a specific circuit configuration ofa radio receiving system according to a fifth embodiment of the presentinvention;

FIGS. 19A and 19B are timing charts for explaining the operation of theradio receiving system according to the fifth embodiment;

FIG. 20 is a circuit diagram showing a specific circuit configuration ofa radio receiving system according to a sixth embodiment of the presentinvention;

FIGS. 21A and 21B are timing charts for explaining the operation of theradio receiving system according to the sixth embodiment;

FIG. 22 is a plot for explaining the principle of the radio receivingsystem according to the sixth embodiment;

FIG. 23 is a block diagram showing the configuration of a receivercircuit of a radio receiving system according to a seventh embodiment ofthe present invention;

FIG. 24 is a block diagram showing the configuration of a receivercircuit of a radio receiving system according to an eighth embodiment ofthe present invention;

FIG. 25 is a block diagram showing the configuration of a receivercircuit of a radio receiving system according to a ninth embodiment ofthe present invention;

FIG. 26A is a schematic representation for explaining how an errorarises in capacitance of a capacitor when it is formed; and

FIG. 26B is a schematic representation for explaining how a capacitorused in a tenth embodiment of the present invention is formed.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Preferred embodiments of the present invention will be described byreference to the accompanying drawings.

(First Embodiment)

FIG. 1 shows a radio receiving system according to a first embodiment ofthe present invention. A radio receiving system 23 comprises a Q-axiscomponent detection/channel-filter section 4, an I signal root Nyquistfilter 20; a Q signal root Nyquist filter 21, and a signaldetection/demodulation section 22. The Q-axis componentdetection/channel filter section 4 further includes a first filter 2which permits passage of only a signal at a frequency band assigned to acommunications system from which the radio receiving system receives asignal; a low-noise automatic gain control HF amplifier 3; asample-and-hold circuit 5; a Hilbert transformer section 6; a firstchannel filter 7; and a second channel filter 2; an N-th channel filter9. The receiver circuit 23 receives an input signal 1 from an antenna, asampling clock signal 10 supplied to the sample-and-hold circuit 5, aclock signal 11 supplied to the Hilbert transformer section 6, a clocksignal 12 supplied to the first channel filter 7, a clock signal 13supplied to the second channel filter 8, a clock signal 14 supplied tothe N-th channel filter 9, a reference clock signal 16, and a clockcontrol signal 17.

The operation of the receiver circuit will be described by reference toFIG. 1. The present invention is in principle based on the patentapplications (Japanese Patent Application Laid-open No. Hei-9-266452“Receiving System” and Japanese Patent Application No. Hei-9-28271“Receiving System,” both being filed by the applicant of the presentpatent application), and hence additional features newly provided forthe receiver circuit in the present invention and the principle requiredby the features will be described.

Turning to FIG. 1, the input signal 1 received from the antenna passesthrough the first filter 2 which passage of only a signal at a frequencyband assigned to a communications system from which the radio receivingsystem receives a signal. After having been amplified by the low-noiseHF amplifier 3, the input signal flows through the sample-and-holdcircuit 5, the Hilbert transformer section 6, and a group of channelfilters including the first channel filter 7 to the N-th channel filter9. The Hilbert transformer section 6 and the group of channel filtersincluding the first channel filter 7 to the N-th channel filter 9receive various types of clock signals 10, 11, 12, 13, and 14 generatedby a clock signal generator comprising a clock signal shaping/controlsection 15 and serve as a channel filter.

An output from the orthogonally-modulated waveform detection/channelfilter section 4 is an orthogonally-modulated waveform detection outputwhich is supplied as an I signal output 18 to the I signal root Nyquistfilter 20 and as a Q signal output 19 to the Q signal root Nyquistfilter 21. After having been shaped so as to have the Nyquistcharacteristics, these signals are demodulated into a base band signal25 by means of the signal detection/demodulation section 22.

FIG. 11 shows specific examples of an I-axis component separationcircuit 61 and a Q-axis component separation circuit 62 of the Hilberttransformer section 6 shown in FIG. 1, and FIG. 12 shows operations offrequency dividers.

Turning to FIG. 11, the sample-and-hold signal output from thesample-and-hold circuit 5 is supplied to a switch SW21 and a switchSW31. An inverting amplifier U1 serves as a shunt feedback amplifier bymeans of negative feedback from a capacitor C3. When the switches SW21and SW22 are in a state such as that shown in FIG. 11, the output fromthe sample-and-hold circuit 5 is specified by a terminal voltagedetermined by electric charges stored in the capacitor C1.

When the switches S21 and S22 are inverted as time t0, the capacitorC2—which is connected to an output terminal of the inverting amplifierU1 and which is charged by the voltage output from the invertingamplifier so far—is connected to an input terminal of the invertingamplifier U1 by means of the switch SW22. In a case where the capacitorC2 is the same in capacitance as the capacitor C3, the potential of theoutput voltage of the inverting amplifier U1 is held in the samepotential. In the meantime, the switch SW21 connects the output terminalof the sample-and-hold circuit 5 to the capacitor C1, whereby a voltagecorresponding to a new sampling value is stored in the capacitor C1.

When the switches SW21 and SW22 again return, at time t1, to the statesuch as that shown in FIG. 11, the output terminal of the capacitor C1in which the voltage output from the sample-and-hold circuit 5 is storedis connected to the inverting amplifier U1. If the capacitors C1 and C3have the same capacitance, a voltage corresponding to a new samplingvalue is produced at the output terminal of the inverting amplifier U1.In short, the inverting amplifier U1 serves as a buffer amplifier—havingthe same polarity as that of the sample-and-hold circuit 5—with respectto the output from the sample-and-hold circuit 5.

In contrast, an inverting amplifier U2 serves as a shunt feedbackamplifier which receives negative feedback from a capacitor C6. Whenswitches SW32 and SW33 are held in a state such as that shown in FIG.11, the output from the sample-and-hold circuit is specified by aterminal voltage determined by electric charges stored in the capacitorC5. At this time, a capacitor C4 is charged by the voltage output fromthe sample-and-hold circuit 5.

When the switches S31 and S32 are inverted as time t0, the capacitorC5—which is connected to an input terminal of the inverting amplifier U2and which dominates the output voltage of the sample-and-hold circuit 5so far—is connected to an output terminal of the inverting amplifier U2by means of the switch SW33. At the same time, the switch SW31 isconnected to ground, and the switch SW32 is connected to the inputterminal of the inverting amplifier U2. Accordingly, if the capacitorsC4 and C6 have the same capacitance, a voltage corresponding to thesampling value output from the sample-and-hold circuit 5 is produced atthe output terminal of the inverting amplifier U2. Further, at the sametime, since the capacitor C5 is connected to the output terminal of theinverting amplifier U2 by means of the switch SW33, the output from theinverting amplifier U2 is stored in the capacitor C5.

When the switches SW32 and SW33 again return, at time t1, to the statesuch as that shown in FIG. 11, the output terminal of the capacitor C1is connected to the input terminal of the inverting amplifier U1, thusholding the potential of the output from the sample-and-hold circuitstill further. In short, the inverting amplifier U2 serves as anamplifier which inverts the polarity of the output from thesample-and-hold circuit 5.

A D-type flip-flop U3 receives as an input the sampling clock signal 10,and an inverted output Q is fed back to an input terminal D of theflip-flop, thus constituting a frequency divider. Similarly, a flip-flopU4 also constitutes a frequency divider. As a result of these flip-flopsU3 and U4 being cascaded, a frequency passing through these frequencydividers is divided by a factor of four.

Operations of the frequency dividers will now be described through useof examples of signal operations provided in the operation timing shownin FIG. 12. The sampling clock signal arrives at the frequency dividersat even time intervals: that is, t1, t2, t3, t4, t5, t6, t7, t8, t9, . .. . The waveform of the sampling clock signal is a rectangular waveformhaving a duty ratio of about 50% as mentioned previously. Upon receiptof such a sampling clock signal, the flip-flop U3 produces a logical 1as an output Q signal at odd timings t1, t3, t5, t7, . . . Upon receiptof the outputs from the flip-flip U3, the flip-flop U4 produces alogical 1 as an output Q signal at timing t1, t2, t5, t6, t9, . . .

In order to separate the orthogonally-modulated signal is divided intotwo signal components in a phase space, all that you have to do is tosubject the orthogonally-modulated signal to phase discrimination at thesame frequency. An orthogonally-modulated waveform detection operationis equivalent to sampling of a signal with an offset of only π/2. Inorder to produce the phase-discriminated signals from a series of valuessampled through one sampling operation, the orthogonally-modulatedsignal is subjected to multiplication and discrimination through use ofcosine and sine functions.

It is though that if the received signal is sampled at the limit of thesampling frequency, the thus-sampled values correspond to a samplingclock signal having a time interval of π/2 as shown in FIG. 12A. Inshort, the highest frequency component is sampled through use of fourpulses of the sampling clock signal shown in FIG. 12A. At this time, itis only required to simultaneously sample rectangular components whilethe cosine and sine functions for the purpose of extracting quadraturecomponents are set to a frequency corresponding to the highestfrequency.

In short, in a case where a sine waveform whose frequency is a quarterof a cyclic frequency of the sampling clock signal “a” is sampledthrough use of the sampling clock signal, +1, −1 of the cosine functionare sampled at positions such as those shown in FIG. 12D, and +1, −1 ofthe sine function are sampled at positions which are delayed only π/2corresponding to one sample relative to those of the cosine function ina manner as shown in FIG. 12E.

Accordingly, I-axis components are obtained by inverting the polarity ofthe sample signal at the positions shown in FIG. 12D, and Q-axiscomponents are obtained by inverting the polarity of the sample signalat the positions shown in FIG. 12E.

Through the foregoing operations, a sample output signal which isequivalent to that obtained by sampling the orthogonally-modulatedsignal can be produced from the sample value obtained through a singlechannel of sampling operations. If the I-axis multiplication factorshown in FIG. 12D and the Q-axis multiplication factor shown in FIG. 12Eare divided into group for every segment of the same polarity in orderto cause a circuit to perform the foregoing sampling operations, it isapparent from FIG. 12F that a pair is formed every two sampling periods.

Since the I-axis quadrature component and the Q-axis quadraturecomponent are alternately output, they can be illustrated in a mannersuch as that shown in FIG. 12G. It is obvious that an output from theforegoing cascaded flip-flops is effective in managing such variationsin the state of output through use of a circuit.

The basic principle and configuration of the channel filter includingthe complex coefficient filter are described in the patent applications(Japanese Patent Application Laid-open No. Hei-9-266452 “ReceivingSystem” and Japanese Patent Application No. Hei-9-28271 “ReceivingSystem,” both being filed by the applicant of the present patentapplication). Hence, only the additional features newly provided for thepresent invention will be explained in detail.

FIG. 2 shows the basic configuration of the channel filter including thecomplex coefficient filter. The drawing is composed of FIGS. 1, 11, 19,21, 24, 29, and 32A of the specification of the foregoing patentapplication (Japanese Patent Application Laid-open No. Hei-9-266452“Receiving System”).

The theory of frequency characteristics of a three-stage complexcoefficient filter—which is the heart of the channel filter shown inFIG. 2—will now be described. FIG. 3 is a view for explaining thefrequency characteristics of a single-stage complex coefficient filter.Provided that a signal vector which is rotated and which has amplitudeAo is located at point P0 at time t0 and that a sampling operation isperformed at every time τ, the next signal vector P+1 is located at aposition shifted from the point P0 by only the product of angularfrequency ω and time τ. In contrast, provided that the amount ofrotation of a vector for the purpose of eliminating adjacent waveformsis θ, the vector P0 moves to point Pr0 at time t0. An output from thecomplex coefficient filter corresponds to the sum of the vector P+1 andthe vector Pr0. I-axis components and Q-axis components which are thequadrature components of the orthogonally-modulated signal arerespectively expressed as follows:

Ir(nT)=I(τ)+Ir=Ao(cos ωτ+cos θ)

Qr(nT)=Q(τ)+Qr=jAo(sin ωτ−sin θ)

The frequency characteristics of the resultant vector are expressed byan envelope of the signal vector, i.e., by input power. Morespecifically, $\begin{matrix}{{Power} = \quad {{{{Ir}({nT})}}^{2} + {{{Qr}({nT})}}^{2}}} \\{= \quad {{{{Ao}\left( {{\cos \quad {\omega\tau}} + {\cos \quad \theta}} \right)}}^{2} + {{{jAo}\left( {{\sin \quad {\omega\tau}} - {\sin \quad \theta}} \right)}}^{2}}} \\{= \quad {{{Ao2}\left\{ {{\cos \quad 2{\omega\tau}} + {2\cos \quad \omega \quad \tau \quad \cos \quad \theta} + {\cos \quad 2\quad \theta}} \right\}} +}} \\{\quad {{Ao2}\left\{ {{\sin \quad 2{\omega\tau}} + {2\sin \quad {\omega\tau}\quad \sin \quad \theta} + {\sin \quad 2\theta}} \right\}}} \\{= \quad {{{Ao2}\left\{ {{\cos \quad 2\quad {\omega\tau}} + {\sin \quad 2{\omega\tau}} + {\cos \quad 2\theta} + {\sin \quad 2\quad \theta}} \right\}} +}} \\{\quad {{Ao2}\left\{ {{\cos \quad \omega \quad \tau \quad \cos \quad \theta} - {2\quad \sin \quad {\omega\tau}\quad \sin \quad \theta}} \right\}}} \\{= \quad {2{Ao2}{\left\{ {1 + {\cos \left( {{\omega\tau} + \theta} \right)}} \right\}.}}}\end{matrix}$

The foregoing expression represents that power corresponds to thefrequency characteristics of the vector which has the angular frequencyω and which includes sampling interval τ and the amount of rotationalphase shift θ. Provided that Ao=1, that the gain of the complexcoefficient filter is 1, and that sampling interval τ is 1, a frequencyfunction which uses ω as a variable is shown in FIGS. 4A to 4D.

FIG. 4A shows the characteristics of a first-stage complex coefficientfilter block shown in FIG. 2, taking the rotation of phase angle of thefilter as −π/4. FIG. 4B shows the characteristics expressed in decibels.FIG. 4C shows the characteristics of the complex coefficient filtersprovided in the first, second, and third stages. These filters rotate aphase angle by −π/4, −2 π/4, and −3 π/4, respectively. FIG. 4D shows thesynthetic characteristics of the filters which are shown in FIG. 4C andare cascaded into three stages. The center frequency of one of theadjacent waveforms is attenuated to −125 dB or more. There is also anattenuation of −25 dB in the boundary region between the adjacentwaveforms.

As described in the patent application (Japanese Patent ApplicationLaid-open No. Hei-9-266452 “Receiving System”), the foregoingcharacteristics ensure a filter frequency range of four channels at eachof the upper and lower sides of the sampling frequency. Further, a zerofrequency of three-stage comb filters—which eliminate waveforms otherthan a desired waveform through use of an offset frequency correspondingto the baseband frequency at the time of a frequency conversionoperation—is placed at the center frequency between adjacent waveforms.More specifically, the channel filter is characterized by being able toensure four channels between image frequencies by means of the samplingfrequency.

The channel filter is followed by equalizers. The amount of phasecompensation required by the equalizer will be theoretically explainedby reference to FIGS. 5 and 6. FIG. 5A shows an explanation about thecomplex coefficient filter I which eliminates the first adjacentwaveform. As shown in FIG. 5A, provided that an adjacent waveform in anegative frequency band (−ω0) is placed at position vector Po whosephase is zero, the phase of the waveform shifts clockwise to positionvector P−1 by only π/8 after one sampling clock. At this time, in orderto produce, from position vector Po, vector Pro which cancels vectorposition P−1, position vector Po is rotated counterclockwise by only 7π/8. FIG. 5A will be mathematically expressed by the following equation.It is obvious that the vector Pro is produced by multiplying the vectorP0 at time to by cos θ and sin θ. $\begin{matrix}{{{Ir}({nT})} = \quad {{Ao}_{—}E\quad {\cos \left( {{{- \omega}\quad {o({t0})}} + \theta} \right)}}} \\{= \quad {{{Ao}_{—}E\quad {\cos \left( {{- \omega}\quad {o({t0})}} \right)} \times \cos \quad \theta} - {{Ao}_{—}E\quad {\sin \left( {{- \omega}\quad {o({to})}} \right)} \times}}} \\{\quad {\sin \quad \theta}} \\{= \quad {{{{I0}({to})} \times \cos \quad \theta} - {{{Q0}({to})} \times \sin \quad \theta}}} \\{{{Qr}({nT})} = \quad {{Ao}_{—}E\quad {\sin \left( {{{- \omega}\quad {o({t0})}} + \theta} \right)}}} \\{= \quad {{{Ao}_{—}E\quad {\sin \left( {{- \omega}\quad {o({t0})}} \right)} \times \cos \quad \theta} - {{Ao}_{—}E\quad {\cos \left( {{- \omega}\quad {o({to})}} \right)} \times}}} \\{\quad {\sin \quad \theta}} \\{= \quad {{{{Q0}({to})}\cos \quad \theta} + {{{I0}({to})} \times \sin \quad {\theta.}}}}\end{matrix}$

Likewise, FIGS. 5B and 5C show phase rotation angles for the purpose ofeliminating adjacent waveforms in the next and third adjacent channels.

A phase rotation angle used for eliminating an adjacent waveform in thenext adjacent channel is 5 π/8, and a phase rotation angle used foreliminating an adjacent waveform in the third adjacent channel is 3 π/8.

As mentioned previously, the phase rotation signifies that the phasecharacteristics are distorted. FIG. 6 shows phase distortion caused byeach of the complex coefficient filters I to III. The phasecharacteristics of the signal after the signal has passed through thethree-stage filter are designated by reference symbol AB shown in FIG.6. A phase offset value is 15π/16 represented by point B.

An explanation will now be given of the foregoing descriptions. Anull-point of the complex coefficient filter I is set to the centerfrequency −fb of the adjacent waveform; a null-point of the complexcoefficient filter II is set to the center frequency −3fb of theadjacent waveform; and a null-point of the complex coefficient filterIII is set to the center frequency −5fb of the adjacent waveform. Bymeans of the 16-times oversampling frequency, a variation in phase ofthe desired waveform corresponding to the period of one sampling time isπ/8. A phase difference of −π/8 arises in the adjacent waveform havingthe center frequency −fb during the period of one sampling time. Inorder to cancel a currently-sampled signal vector by rotation of apreviously-sampled signal vector, the previously-sampled signal vectoris rotated by only 7π/8. At this time, the desired waveform has a phasedifference of 6π/8 relative to the canceled signal vector and henceremains as a vector of 2 sin(π/8). In short, assuming that with anI-axis component of a value sampled at time t0 is expressed as Io and anQ-axis component of the same is expressed as Qo, rotation of each of theposition vectors at time t1 is represented by

I-axis Position Vector=Io×cos 7π/8−Qo×sin 7π/8

and

Q-axis Position Vector=Io×sin 7π/8+Qo×cos 7π/8,

respectively.

With regard to a resultant vector which is a sum of a signal vectorsampled at time t1 and a vector obtained by rotating the signal vectorsampled at the preceding sampling time, an I-axis component output fromthe complex coefficient filter I is expressed as I1 and an Q-axiscomponent output from the same is expressed as Q1, each beingrepresented by

I 1=Io(t=to+ts)+Io(t=to)×cos 7π/8−Qo(t=to)×sin 7π/8  (Eq. 1)

and

Q 1=Qo(t=to+ts)+Io(t=to)×sin 7π/8+Qo(t=to)×cos 7π/8  (Eq. 2),

where t and t0 represent time, and ts represents a sampling timeinterval.

Similarly, the complex coefficient filter II for the purpose ofeliminating the adjacent waveform in the next adjacent channel rotatesthe position vector by 5π/8, and the complex coefficient filter III forthe purpose of eliminating the adjacent waveform in the adjacent channelafter the next channel rotates the position vector by 3π/8. An I-axiscomponent output from the complex coefficient filter II is expressed asI2; and a Q-axis component output from the same as Q2, each of which isrepresented by

I 2=I 1(t=to+ts)+I 1(t=to)×cos 5π/8−Q 1(t=to)×sin 5π/8  (Eq.3),

and

Q 2=Q 1(t=to+ts)+I 1(t=to)×sin 5π/8+Q 1(t=to)×cos 5π/8  (Eq.4).

An I-axis component output from the complex coefficient filter III isexpressed as I3, and a Q-axis component output from the same as Q3, eachof which is represented by

I 3=I 2(t=to+ts)+I 2(t=to)×cos 3π/8−Q 2(t=to)×sin 3π/8  (Eq.5)

and

Q 3=Q 2(t=to+ts)+I 2(t=to)×sin 3π/8+Q 2(t=to)×cos 3π/8  (Eq.6).

With regard to FIG. 5B, taking the center frequency of the desiredwaveform as +ωo and the center frequencies of respective adjacentwaveforms in the lower adjacent three channels as −ωo, −3ωo, and −5ωo,the phase characteristics P of the complex coefficient filters I, II,and III which eliminate the adjacent waveforms are respectivelyrepresented by

Complex Coefficient Filter I for eliminating −ωo

P=−πω/16ωo+3π/16

Complex Coefficient Filter II for eliminating −3ωo

P=−πω/16ωo+5π/16

Complex Coefficient Filter III for eliminating −5ωo

P=−πω/16ωo+7π/16

These phase characteristics of three filters are expressed as threerightwardly-declining parallel lines. The sum of phase characteristicsof these three filters is represented by

P=−3πω/16ωo+15π/16

and is designated by line A shown in FIG. 6. An intercept of the line Aatω=0 is represented by point B which corresponds to 15π/16. As aresult, it is obvious that the equalizer is required to cause a phaseshift so as to cancel 15π/16.

It is obvious from FIG. 6 that the phase characteristics of therespective complex coefficient filters I, II, and III are linear.Accordingly, the sum of the characteristics of these filters evidentlybecome linear.

In a case where a plurality of channel filters are connected tandem, thetotal characteristics of the channel filters are represented by thefollowing equation, and it is obvious that the total characteristics arerepresented by a linear function with regard to a frequency. Morespecifically, the phase characteristics of the first-stage channelfilter are expressed as $\begin{matrix}{{P1} = \quad {\left\{ {{{{- {\pi\omega}}/16}\omega \quad o} + {3{\pi/16}}} \right\} + \left\{ {{{{- {\pi\omega}}/16}\omega \quad o} + {5{\pi/16}}} \right\} +}} \\{\quad \left\{ {{{{- {\pi\omega}}/16}\omega \quad o} + {7{\pi/16}}} \right\}} \\{= \quad {{{- 3}{{\pi\omega}/16}\omega \quad o} + {15{\pi/16.}}}}\end{matrix}$

Assuming that the channel filter in the second stage diminishes thesampling frequency of the sampling clock signal to a quarter of itsoriginal frequency, the frequency of the desired waveform on the secondstage becomes ωo/4, and hence the phase characteristics P2 of thesecond-stage channel filter are expressed as $\begin{matrix}{{P2} = \quad {\left\{ {{{- 4}{{\pi\omega}/16}\omega \quad o} + {3{\pi/16}}} \right\} + \left\{ {{{- 4}{{\pi\omega}/16}\omega \quad o} + {5{\pi/16}}} \right\} +}} \\{\quad \left\{ {{{- 4}{{\pi\omega}/16}\omega \quad o} + {7{\pi/16}}} \right\}} \\{= \quad {\left\{ {{{- n}\quad {{\pi\omega}/16}\omega \quad o} + {3{\pi/16}}} \right\} + \left\{ {{{- n}\quad {{\pi\omega}/16}\omega \quad o} + {5{\pi/16}}} \right\} +}} \\{\quad \left\{ {{{- n}\quad {{\pi\omega}/16}\omega \quad o} + {7{\pi/16}}} \right\}} \\{= \quad {{{- 3}n\quad {{\pi\omega}/16}\omega \quad o} + {15{\pi/16.}}}}\end{matrix}$

The phase characteristics Pn of the n-th channel filter are expressed as$\begin{matrix}{{Pn} = \quad {\left\{ {{{- 4}\quad {{\pi\omega}/64}\omega \quad o} + {3\quad {\pi/16}}} \right\} + \left\{ {{{- 4}n\quad {{\pi\omega}/64}\omega \quad o} + {5\quad {\pi/16}}} \right\} +}} \\{\quad \left\{ {{{- 4}\quad {{\pi\omega}/64}\omega \quad o} + {7\quad {\pi/16}}} \right\}} \\{= \quad {\left\{ {{{- n}\quad {{\pi\omega}/16}\omega \quad o} + {3\quad {\pi/16}}} \right\} + \left\{ {{{- n}\quad {{\pi\omega}/16}\omega \quad o} + {5\quad {\pi/16}}} \right\} +}} \\{\quad \left\{ {{{- n}\quad {{\pi\omega}/16}\omega \quad o} + {7\quad {\pi/16}}} \right\}} \\{= \quad {{{- 3}n\quad {{\pi\omega}/16}\omega \quad o} + {15\quad {\pi/16.}}}}\end{matrix}$

Accordingly, the total characteristics Ptotal of the n-th stage channelfilter are represented by $\begin{matrix}{{Ptotal} = {{\Sigma \quad n\quad i} = {1\left( {{{- 3}i\quad {{\pi\omega}/16}\quad \omega \quad o} + {15\quad {\pi/16}}} \right)}}} \\{= {{{- 3}{n\left( {n - 1} \right)}{{\pi\omega}/32}\quad \omega \quad o} + {15n\quad {\pi/16.}}}}\end{matrix}$

From the foregoing explanations, it is obvious that the phasecharacteristics are represented by a linear function with regard tofrequency ω.

Consequently, it is obvious that the equalizer compensates for thedistortion of phase characteristics by rotating the signal vector by 15nπ/16.

<Channel Filters Connected Tandem>

In a case where a plurality of channel filters, each including theforegoing complex coefficient filter, are connected tandem, it isevident that diminishing of the sampling frequency to a quarter of itsoriginal frequency enables realization of a very efficient filter havingsuperior symmetry in terms of frequency characteristics.

FIGS. 7A to 7C show the theoretical characteristics of the channelfilters when two channel filters are connected tandem.

FIG. 7A shows the frequency characteristics of the channel filter shownin FIG. 2 when it is activated at an oversampling frequency which is 64times as wide as the frequency bandwidth of the channel filter. FIG. 7Bshows the frequency characteristics of the channel filter when it isactivated in the same manner as that shown in FIG. 4B at an oversamplingfrequency which is 16 times as wide as the frequency bandwidth of thechannel. FIG. 7C shows the total characteristics of these two filtershaving the foregoing characteristics. It is obvious from FIG. 7C that 16channels are contained between the sampling waveforms. The signalpassing through the boundary region between the next adjacent waveformand the adjacent waveform after the next is attenuated to −30 dB, andthe signal passing through the boundary region between the seventhadjacent waveform and the eighth adjacent waveform is attenuated to −60dB or more. As a matter of course, a null point arises in the filter atthe center frequency of the adjacent waveform, and there arises anattenuation of −125 dB or more.

<Elimination of a Frequency Offset>

FIG. 8 shows a specific example in which the channel filters, eachincluding a complex coefficient filter, are connected tandem. In FIG. 8,an input signal enters the band pass filter, and a desired bandcomponent is output. The thus-output signal is sampled at the frequencyof the sampling clock signal in the sample-and-hold circuit. An outputfrom the sample-and-hold circuit is separated into quadrature componentsby the Hilbert transformer section. The quadrature components aredelivered to the channel filters 1 and 2 connected tandem.

In the channel filter 1, the quadrature components pass through thecomplex coefficient filters I, II, and III, thus permitting passage of adesired band component. Errors in rotation of phase of the quadraturecomponents caused by the complex coefficient filters are compensated bythe equalizers provided in a subsequent stage. The quadrature componentspass through averaging circuits which act as low pass filters foreliminating repeated noise in a high frequency range, entering frequencydividers or sample-and-hold circuits which convert the samplingfrequency to a lower sampling frequency required by the channel filter 2provided in a subsequent stage. Subsequently, the offset frequency ofthe quadrature components is changed. The offset frequency of thequadrature components must be maintained at the same frequency evenafter the components having finished undergoing the frequency dividingoperation.

As shown in FIG. 7A, a desired waveform is assigned a frequency offsetwhich is half the bandwidth of the desired waveform or is a quarter ofthe pass bandwidth of the channel filter 1. If the sampling frequency isdiminished and supplied to the channel filter 2, the frequency offset isreduced to a quarter of the pass bandwidth determined by the channelfilter 2. As a result, a discrepancy arises between the centerfrequencies of the pass bandwidths shown in FIGS. 7A and 7B. To preventsuch a discrepancy, a frequency offset circuit is provided at the end ofthe channel filter 1.

The outputs from the channel filter 1 are supplied to the channel filter2. The channel filter 2 is the same in operation and configuration asthe channel filter 1, except that the sampling frequency used in thechannel filter 2 is different from that used in the channel filter 1.Filtering effect such as that shown in FIG. 7B is obtained.

Accordingly, the multiband receiving filter shown in FIG. 8 exhibits thetotal characteristic such as that shown in FIG. 7C. The frequencyoffsets added to the quadrature components so far are eliminated by afrequency offset circuit provided at the end of the channel filter 2,and I and Q signals are output as complete baseband signals.

From the foregoing descriptions, the theory underlying the firstembodiment of the present invention becomes obvious. The firstembodiment will further be described. FIG. 10D shows the influence of anaperture effect due to a sampling operation, i.e., the spectralcharacteristics of the sampling operation {sin(πf/fs)}/(πf/fs), on thesynthetic characteristics of the three-stage filters shown in FIG. 4C orthe characteristics of the multiband receiving filter shown in FIG. 7C.

To prevent the influence of the aperture effect, the frequencycharacteristics of the first filter (i.e., a band pass filter) 2—for thepurpose of permitting passage of only a signal at a frequency bandassigned to a communications system from which the radio receivingsystem according to the first embodiment receives a signal—are set so asto become opposite to the aperture effect due to a sampling operation ina manner such as that shown in FIG. 15. The band pass filter having suchcharacteristics can be readily realized through use of a SAW (surfaceacoustic wave) filter.

(Second Embodiment)

As mentioned previously, in order to highly accurately perform asampling operation, previous and subsequent stages of the samplingcircuit must have high speed characteristics required to ensure over aconsiderably wide range the speed performance of the sampling circuitfor the purpose of preventing the aperture effect. For example, in orderto sample a signal of 1 MHz having a carrier wave of 100 MHz is sampledwith an accuracy of 8 bits, the aperture time must be reduced to 13.67ps or less. In order to provide such high speed characteristic to thesampling circuit, the capacitance of a capacitor for holding purposemust be reduced to several pF. As a result, the hold circuit includingthe hold capacitor is provided with high speed characteristics, and acircuit having a bandwidth of 100 MHz is used for extracting a signalhaving a bandwidth of 1 MHz.

In other words, the bandwidth of the hold circuit becomes considerablywider than that of the received signal. In spite of a band pass filterbeing provided in a previous stage for the purpose of limiting thereceiving signal to a predetermined bandwidth, the bandwidth of acircuit provided in a subsequent stage must become considerably widerthan that of the band pass filter. The amount of thermal noise caused bythe hold circuit becomes much larger than that caused in the existingreceiver circuit, which in turn accounts for an increase in the errorrate. Even an existing circuit used with an IF signal such as that shownin FIG. 9 has a disadvantage of a wide bandwidth of circuits followingthe hold circuit, as well as of an increase in thermal noise. A radioreceiving system according to a second embodiment of the presentinvention is one which forms a channel filter by quantizing a receivedsignal through sampling and by subjecting the thus-quantized signal todigitized signal processing operation. The radio receiving systemcomprises a sample-and-hold circuit for sampling and holding an outputfrom a band pass filter which compensates for the aperture effect due toa sampling operation, and an integrating circuit having the function ofintegrating the received signal during a sampling period of thesample-and-hold circuit.

FIG. 13 is a circuit diagram for explaining the radio receiving systemaccording to a second embodiment of the present invention. As shown inFIG. 13, an integrating circuit having an integrating function is addedto the sample-and-hold circuit 5 shown in FIG. 1 used for explaining thefirst embodiment of the present invention.

In FIG. 13, the radio receiving system according to the secondembodiment comprises a sample-and-hold circuit A and an integratingcircuit B. An input signal is supplied to the sample-and-hold circuit Aand the integrating circuit B. In the sample-and-hold circuit A, aninput signal 101 is divided into two signal components by a divider 102.The thus-divided two signals are amplified respectively by couplingcapacitors 103 and 104 and are fed to amplifying transistors 105 and106, respectively. Outputs from the amplifying transistors 105 and 106are delivered to sample-and-hold switching transistors 113, 114 and 115,116. Gate circuits 119 and 120 for the purpose of controlling thesample-and-hold switching transistors 113, 114 and 115, 116 arecontrolled by a sampling pulse amplifier 118 which receives as an inputa sampling pulse signal 117.

Output terminals of the sample-and-hold transistors 113 and 114 areconnected together by means of an output line 121 connected to a holdcapacitor 122. Output terminals of the switching transistors 115 and 116is grounded, thus permitting flow of the input signal during a holdperiod. A terminal voltage of the hold capacitor 122 is increased by abuffer amplifier 124 and is output from a sample-and-hold output signal125.

The amplifying transistors 105 and 106 are connected respectively tocurrent mirror transistors 107 and 108 by way of buffer resistors 111and 112. The current mirror transistors 107 and 108 receive a constantcurrent from constant current power supplies 109 and 110. The samplingpulse signal 117 is assigned a waveform operation such as that shown inFIG. 14B.

In the integrating circuit B, the input signal 101 is supplied to anattenuator 1100, and the amount of attenuation of the attenuator 1100 iscontrolled by means of an attenuation control signal 1101. An outputfrom the attenuator 1100 is divided into two signal components by adivider 1102. The thus-divided two signals are amplified respectively bycoupling capacitors 1103 and 1104 and are fed to amplifying transistors1105 and 1106, respectively. Outputs from the amplifying transistors1105 and 1106 are delivered to sample-and-hold switching transistors1113, 1114 and 1115, 1116.

Gate circuits 1119 and 1120 for the purpose of controlling thesample-and-hold switching transistors 1113, 1114 and 1115, 1116 arecontrolled by a sampling pulse amplifier 1118 which receives as an inputan integration control signal 1117. Output terminals of thesample-and-hold transistors 1113 and 1114 are connected together bymeans of an output line 1121 connected to a hold capacitor 1122.

Output terminals of the switching transistors 1115 and 1116 is grounded,thus permitting flow of the input signal during a hold period. Aterminal voltage of the hold capacitor 1122 is increased by the bufferamplifier 124 and is output from the sample-and-hold output signal 125.The amplifying transistors 1105 and 1106 are connected respectively tocurrent mirror transistors 1107 and 1108 by way of buffer resistors 1111and 1112. The current mirror transistors 1107 and 1108 receive aconstant current from constant current power supplies 1109 and 1110. Theintegration control signal 117 is assigned a waveform operation such asthat shown in FIG. 14C.

The sample-and-hold circuit A is different from the integrating circuitB in two points: First, they use different control signals. Second, thecoupling capacitors 103 and 104 are connected to the drains of thecorresponding amplifying transistors 105 and 106 and are directlyconnected to a bridge circuit comprising the transistors 113, 114, 115,and 116. In contrast, the coupling capacitors 1103 and 1104 areconnected to the gates of the corresponding amplifying transistors 1105and 1106 and are connected to a bridge circuit comprising thetransistors 1113, 1114, 1115, and 1116 by way of the drains of theamplifying transistors 1105 and 1106.

As a result, with the circuit configuration formed from the couplingcapacitors 103 and 104 and the amplifying transistors 105 and 106, theinput signal 101 is supplied as a voltage signal to the bridge circuitcomprising the transistors 113, 114, 115, and 116. The signals inputfrom the coupling capacitors 1103 and 1104 are converted from voltagesignals to current signals by means of the amplifying transistors 1105and 1106, and the thus-converted current signals are supplied to thebridge circuit comprising the transistors 1113, 1114, 1115, and 1116.

FIGS. 14A to 14C show the theory of operation of the radio receivingsystem according to the second embodiment of the present invention shownin FIG. 13. As shown in FIGS. 14A to 14C, the input signal is sampled att1, t2, t3, t4, and t5. The sample-and-hold circuit A changes to anintegrating state from a tracking state—in which the circuit is heldimmediately before the sampling timing—at times t1, t2, t3, t4, and t5.

As shown in FIGS. 14A to 14C, the tracking state is represented asintervals from t1p to t1, from t2p to t2, from t3p to t3, from t4p tot4, and from t5p to t5. The input signal is integrated during a periodof Δt between t1 and t1s, that between t2 and t2s, that between t3 andt3s, that between t4 and t4s, and that between t5 and t5s.

The circuit operates on the basis of the same principle throughoutoperations, and hence the operation of the circuit at time t1 will beexplained as an example. The sample-and-hold circuit A commences atracking operation at time t1p and enters a hold state at time t1. Thevoltage of the input signal 101 at time t1 is P1, and a potential P′1 isreceived from a hold capacitor CH by way of the sample-and-hold circuit.Subsequently, the input signal 101 is converted into an electric currentby means of the transistors 1105 and 1106 of the integrating circuit.During a period from t1 to t1p, the transistors 1113 and 1114 arebrought into conduction, thus storing an electric current into the holdcapacitor 122. In other words, the signal input to the integratingcircuit from t1 to t1p is integrated, and the thus-obtained integratedvalue is superimposed on the potential P′1, thus producing a potentialP″1. When the sample-and-hold operation and the integrating operationare performed, an electric potential expressed by Eq.7 is stored in thehold capacitor 122. When the integrating operation is performed, anelectric potential expressed by Eq.8 is stored in the hold capacitor122.

With a view to simplifying calculation, the input signal is expressed asa simple sine wave (i.e., having an amplitude of 1). Further, therelationship between the output from the sample-and-hold circuit A andthe output from the integrating circuit B is controlled by theattenuator 1100 which controls the amount of attenuation through use ofthe attenuation control signal 1101.

(Terminal Voltage of Hold Capacitor)=cos 2

πfct(t=t 1)+_(—) çt

1+t 1Δt cos 2πfctdt=cos 2

πfct(t=t 1)+[(1/2πfc)sin 2

πfct]t 1 t 1+Δt=cos 2

πfct 1+(1/2πfc){sin 2

πfc(t 1+Δt)−sin 2

πfcΔt}  (Eq.7)

Assuming that a sampling point t1 is placed at the center of Δt and Δtis placed at 2 Δτ in order to simplify calculation, the foregoingexpression will become

(Terminal Voltage of Hold Capacitor)=cos 2

πfct 1+(1/2πfc){sin 2

πfc(t 1+Δτ)−sin 2

πfc(t 1−Δτ)}=cos 2

πfct 1+(1/2πfc){sin 2

πfct 1 cos 2πfcΔτ+cos 2

πfct 1 sin 2πfcΔτ−sin 2

πfct 1 cos 2πfcΔτ+cos 2

πfct 1 sin 2πfcΔτ}=cos 2

πfct 1+(1/πfc)cos 2

πfct 1 sin 2πfcΔτ=cos 2

πfct 1{1+(1/πfc)sin 2

πfcΔτ}  (Eq.8)

In short, it is obvious that a variation of sin 2πfc Δτ is added to anoutput of cos 2πfct1 resulting from sampling of a voltage, by theneglect of the fact that an amplitude factor of 1/πfc vulnerable to afrequency is multiplied by the output of cos 2πfct1. From thisdescription, it can be decided that the input signal is not distortedwithin the range in which Δτ does not cause a quantization error in theinput signal.

Since the actually input signal includes various interference waves ornoise present in a limited bandwidth, consideration must be given todisturbance corresponding to a ratio of the carrier frequency to thebandwidth rather than to a simple sine wave. The value of disturbance isusually considered to be 0.1 or less, and hence the value can besubstantially calculated by the foregoing equations. Accordingly, if aquantization accuracy is eight bits, the width of Δτ is set to 1/256 orless of one cycle of a data rate.

Assuming that the bandwidth of the first filter 2—that permits passageof only a signal a frequency band assigned to a communications systemfrom which the radio receiving system receives a signal—is 1 MHz, Δτ canbe extended to 1/256 of 1 μs, i.e., 4 ns or less.

The integration of the received signal has the effect of being able tointegrating the energy of the desired waveform signal. Particularly,even in a case where a weak radio wave is received and a desiredwaveform signal is buried in thermal noise in a circuit, a samplingoperation being performed at an ordinary voltage enables power to beproduced from an input signal only during a period over which theaperture effect arises. However, the radio receiving system according tothe present embodiment integrates the received signal while the periodof time during which the energy is integrated is extended, thus doublingpower. Further, a random signal, such as thermal noise, cancels itselfas a result of integration, enabling an improvement in a signal-to-noiseratio.

Compared with an ordinary sample-and-hold circuit, the sample-and-holdcircuit according to the second embodiment enables the capacitance ofthe hold capacitor to be increased to such an extent as to correspond toa ratio of an aperture time to an integral action time, and hence thehigh frequency impedance of a terminal of the capacitor can be reducedto a considerably small value. Accordingly, mixing of noise andoccurrence of thermal noise can be prevented.

For example, in the foregoing embodiment, the aperture time is 13.67 psor less, and the integral action time is 4 ns or less. A ratio ofaperture time to integral action time is 292.6. Although a holdcapacitor having a capacitance of 0.3 pF or thereabouts is used for theexisting sample-and-hold circuit, the capacitance of the capacitor canbe increased to about 87.8 pF in the second embodiment. In short, anexisting sample-and-hold circuit has a high impedance of 5.30 kilo ohmswith respect to a signal having a bandwidth of 100 MHz, whereas thesample-and-hold circuit according to the present embodiment has a lowimpedance of 18.1 ohms with respect to the same signal. Assuming thatnoise is mixed into the input signal from a noise source having asignal-source impedance of 50 ohms by way of a stray capacitance of 1pF, the hold capacitor having a capacitance of about 0.3 pH causes anattenuation of only −2.56 dB, but the hold capacitor having acapacitance of 87.7 pF causes an attenuation of −39.2 dB, thus reducingnoise to 36 dB or more.

(Third Embodiment)

As represented by Eq.8, the carrier wave frequency fc of the receivedsignal supplied to the sampling circuit and an integral action time Atcorrespond to each other in a straightforward manner. For example, in acase where there is an increase in the received carrier wave frequencyfc, in

(Terminal Voltage of Hold Capacitor)=cos 2πfct 1{1+(1/πfc)sin 2πfcΔτ},

if an integral action period Δτ is not changed with regard to fc in theabove equation, Δτ deviates from a desired value determined byquantization accuracy relative to one cycle of a data rate.

Consequently, in a case where consideration is given to amultiband-compatible radio receiving system: that is, where the samplingfrequency is changed in order to correspond to a change in the carrierwave frequency or bandwidth of a received signal, if an integrationconstant is fixed, an output is driven to saturation or is decreased,impairing the originally expected function of the radio receiving systemaccording to the present invention.

In a case where the sample-and-hold circuit is used while its samplingfrequency is changed, if the sampling period Δt is fixed, the expectedeffect of the second embodiment will be deteriorated as a matter ofcourse.

For example, in a case where there is an increase in the carrier wavefrequency fc of the received signal, the integral action period Δτ ischanged so as to correspond to such an increase in the carrier wavefrequency, the capacitance of CH in Eq.8: that is,

(Terminal Voltage of Hold Capacitor)=(1/2πfcCH)cos 2πfct 1{1+(1/πfc)sin2πfcΔτ},

must be changed, thus decreasing an integral output and hencedeteriorating a signal-to-noise ration.

In order to solve the aforementioned problem, the integrating circuitused in the radio receiving system according to the second embodiment isarranged so as to enable the integral action time to be changed orselected from a plurality of values in the third embodiment.

FIG. 16 is a schematic diagram for explaining a radio receiving systemaccording to the third embodiment of the present invention. A delaydevice shown in FIG. 16 is designed so as to make variable the width ofthe sampling pulse signal 117 to be applied to the sample-and-holdcircuit which is shown in FIG. 13 and which is described for the secondembodiment.

The circuit shown in FIG. 16, as a whole, constitutes a variable delaydevice which receives as an input signal a basic signal 130 and whichproduces the sampling pulse signal 117. The delay instruction signal 132is supplied to the variable delay device in a digital form, and thedelay device manages the degree of delay corresponding to workingconditions and determines the length of the integral action time Δt.

More specifically, the variable delay device comprises adigital-to-analog converter 141 which receives the delay instructionsignal 132, a variable capacity diode 142 which receives an outputvoltage from the digital-to-analog converter 141, a coupling capacitor143 which transmits the electric charges stored in the variable capacitydiode; and a monostable multivibrator 144 which integrates and storesthe electric charge discharged from the capacitor.

Although the variable delay device can change the width of the samplingpulse signal by supplying the delay instruction signal 132 in theforegoing example, the sampling pulse itself may be selected from aplurality of sampling pulse signals.

With the foregoing configuration, the integral action time in thesample-and-hold circuit is changed according to a variation in thefrequency or bandwidth of an input signal, thus realizing the desiredeffectiveness of integration.

(Fourth Embodiment)

In a case where consideration is given to a multiband-compatible radioreceiving system: that is, where the sampling frequency is changed inorder to correspond to a change in the carrier wave frequency orbandwidth of a received signal, there is necessity to change theintegral action time of an integrating circuit added to thesample-and-hold circuit.

If the integration active time is changed while the integral capacity ofthe integrating circuit is fixed, an output is driven to saturation oris decreased, impairing the originally expected function of the radioreceiving system. As represented by Eq.8, the carrier wave frequency fcof the received signal supplied to the sampling circuit and the integralcapacity defined by an integral action time Δt correspond to each otherin a straightforward manner. For example, in a case where there is anincrease in the received carrier wave frequency fc, provided that theintegral action period Δτ is changed so as to correspond to the increasein the frequency of the carrier waveform, the capacitance CH in Eq.8,i.e.,

(Terminal Voltage of Hold Capacitor)=(1/2πfcCH)cos 2πfct 1{1+(1/πfc)sin2πfcΔτ},

A is not changed, an integral output is decreased, resulting indeterioration of a signal-to-noise ratio.

In order to solve the aforementioned problem, the integrating circuitused in the radio receiving system according to the third embodiment isarranged so as to enable the integral action time to be changed orselected from a plurality of values in the fourth embodiment.

FIG. 17 is a schematic diagram for explaining a radio receiving systemaccording to the fourth embodiment of the present invention. FIG. 17shows a variable capacitor added to the hold capacitor 122 of thesample-and-hold circuit 5 which is shown in FIG. 13 and which isdescribed for the third embodiment.

The circuit shown in FIG. 17, as a whole, constitutes a variable holdcapacitor. More specifically, the variable hold capacitor comprises adigital-to-analog converter 151 which receives an integrationinstruction signal 155, a variable capacity diode 152 which receives anoutput voltage from the digital-to-analog converter 151, a couplingcapacitor 153 which transmits the electric charges stored in thevariable capacity diode; and current coupling means 154 which connectsthe digital-to-analog converter 151 to the variable capacity diode 152.

With the foregoing configuration, the integral capacity of thesample-and-hold circuit is changed according to a variation in thefrequency or bandwidth of an input signal, thus realizing the desiredeffectiveness of integration.

(Fifth Embodiment)

Although the method of providing the integrating circuit for reducingnoise has been described in the second embodiment, no weighting isperformed during the integral action time. Consequently, the accuracy ofdetection of a desired signal is decreased. For this reason, theintegral action time is limited to the range defined by Eq.2. In otherwords, the integrating circuit is less effective in reducing lowfrequency noise outside the range.

Since the leading and trailing edges of the sampling pulse signal mustbe made steep, the amplifier which supplies a sampling signal isrequired to have a drive capacity and high speed performance. At thesame time, unnecessary external radiation from the amplifier is apt toincrease.

In order to solve the aforementioned problem, the integrating circuitused in the radio receiving system according to the second embodiment isarranged so as to provide an integral gate function with the waveform ofa Nyquist signal in the fifth embodiment.

FIG. 18 is a schematic diagram for explaining a radio receiving systemaccording to the fifth embodiment of the present invention. The elementsfrom the inputs signal 101 to the sample-and-hold output signal 125 arethe same as those which are shown in FIG. 13 and which have beendescribed for the second embodiment. The sampling pulse signal 117 issupplied to a Nyquist filter 161, and an output 162 from the Nyquistfilter 161 is supplied to the sampling pulse amplifier 118.

FIGS. 19A and 19B are timing charts for explaining operations of thesample-and-hold circuit shown in FIG. 18 through use of waveforms. FIG.19A shows the waveform of the sampling pulse signal 117, and FIG. 19Bshows an output 162 that has passed through the Nyquist filter 161.

A received signal is integrated on the basis of the Nyquist signal shownin FIG. 19B. More specifically, the received signal is integrated as aresult of having passed through a window formed by the waveform of theNyquist signal. A period during which transmission pulse power becomeshalf the original power, i.e., the sample-and-hold period, is taken as τ1, the integral action time is taken as Δt in the manner analogous tothat mentioned previously, and Δt is expressed as 2τ.

In a case where a Nyquist waveform is expressed in the form of a normaldistribution waveform, there will be obtained a relationship such as 3τ1_ . . . Δt. In short, a total of integral action time is approximately1.5 times as longer as that required by the circuit according to thesecond embodiment. Within 67% of the integral action time, the circuitacquires 99% of power of the received signal. Further, substantially allthe pulse frequency components are concentrated at a frequency of 1/2τ,thus preventing higher harmonic waves which would otherwise occur overthe high frequency portion of an ordinary pulse signal.

With the foregoing configuration, an efficient sample-and-holdcircuit—which has an integrating function and which provides a highersignal-to-noise ratio—can be realized.

(Sixth Embodiment)

In any one of the sample-and-hold circuits having the integratingcircuits according to the second through fifth embodiments, theintegrating circuit has integral action time which is shorter than thecycle of a carrier signal of the received signal. For this reason, theintegrating circuit cannot sufficiently eliminate a random signal, suchas thermal noise, from low-frequency components.

A sixth embodiment of the present invention is directed toward solvingthe foregoing problem and is characterized by the feature that the timeconstant of the sample-and-hold circuit of the radio receiving systemaccording to the second or third embodiment is made longer than asampling cycle.

FIG. 20 is a schematic diagram for explaining a radio receiving systemaccording to the sixth embodiment of the present invention. Turning toFIG. 20, the elements from the inputs signal 101 to the sample-and-holdoutput signal 125 are the same as those which are shown in FIG. 13 andwhich have been described for the second embodiment. The systemcomprises a sampling pulse signal 1117, a counter 171 for receiving thesampling pulse signal, a long cycle sample-and-hold signal 172 outputfrom the counter 171, and a control signal 173 for providing the counter171 with an instruction relating to the number of counts. An explanationof a reset signal system will be omitted. Exemplary operations of theradio receiving system are shown in FIGS. 21A and 21B. FIG. 21A showsthe sampling pulse signal 1117, and FIG. 21B shows an output from thecounter 117. The principle of the radio receiving system will beillustrated in FIG. 22.

The sampling pulse signal 1117 shown in FIG. 20 arrives at evenintervals t1n, t2n, t3n, . . . , tn−1n, tnn, tnm3, . . . , tm−1m, tmm, .. . at the counter 171 in the manner as shown in FIG. 21A. Upon receiptof the control signal 173, the counter 171 produces a long cycle pulsewhich causes an integrating operation during a period of n-pulses andholds the signal during a period of m-pulses. The thus-produced longcycle signal is the long cycle sample-and-hold signal 172. Namely, thesample-and-hold signal becomes H at tin and becomes L at tnn and returnsto H at tmm. The sample-and-hold signal repeatedly performs a round ofthese operations. As a result, the cycle of the long cyclesample-and-hold signal 172 is increased to (n+m) times as long as thatof the sampling pulse signal 1117. Such a state of the sample-and-holdsignal is expressed in the form of frequency characteristics.

The curve of characteristics shown in FIG. 22 represents the frequencycharacteristics of an output when the received signal is sampled at asampling frequency fs: that is, generally-called samplingcharacteristics. The frequency characteristics of the output areexpressed by {sin(πf/fs)}/(πf/fs), given that a d.c. range is taken as arelative amplitude of 1.

In general, the input frequency band width of the sampling frequency isdetermined so as to attenuate δa within the range of quantization error.The sixth embodiment of the present invention is characterized by aspectral wave appearing between a frequency position which is four timesas high as the sampling frequency fs and a frequency position which isfive times as high as the same: that is, by use of a spectral waveappearing between 4fs and 5fs. The peak of the spectral wave of samplingcharacteristic appears at a frequency position of about 4.5fs, and alevel of the spectral wave at the peak An+m is about 0.05. It is obviousthat a frequency range fBW within a quantization accuracy of δb can beutilized with reference to the peak of the spectral wave. In otherwords, the relationship between the frequency fc of the received signalto be sampled and the sampling frequency fs is defined as 4.5:1, thecurve of sampling characteristics such as that shown in FIG. 22 isobtained. Although a sampled output is attenuated to 0.05 in the presentexample, noise is integrated at a long cycle which is 4.5 times as longas that of noise. Accordingly, depending on properties of noise, theforegoing sampling operation is very effective in reducing noise.Further, a diminution in sampling frequency contributes to a reductionin the power dissipated by the sample-and-hold circuit and peripheraldevices connected thereto.

(Seventh Embodiment)

In the second through sixth embodiments, a signal componentsubstantially in a d.c. range is sampled. Even if a sample output issubjected to an a.c. coupling operation, it is difficult to remove thed.c. information components extracted by sampling. Consequently, it isdifficult to eliminate unwanted components, such as temperature drift ofthe sampling circuit or a d.c. offset of an input circuit.

A seventh embodiment has been conceived to solve the foregoing problemand provides a radio receiving system in which a channel filter isformed by quantizing a received signal through sampling and bysubjecting the thus-quantized signal to a digitized-signal processingoperation, the system comprising: sampling means which is made of asample-and-hold circuit and which samples the received signal;difference calculation means for calculating a difference between acurrently-sampled signal received from the sample-and-hold circuit and apreviously-sampled signal; and means for calculating a differencebetween the output from the difference calculation means and an outputfrom the band pass filter and inputs the thus-obtained difference to thesample-and-hold circuit.

FIG. 23 is a circuit diagram for explaining a radio receiving systemaccording to a seventh embodiment. In FIG. 23, the following elementsare the same as those shown in FIG. 2: that is, the RF signal 201, theband pass filter 202, the output 203 from the band pass filter 202, thesample-and-hold circuit 204, the sample-and-hold signal 205, thesample-and-hold circuit output 206, the Hilbert transformer circuit 207,the clock signal 298 for Hilbert transformation purpose, the initialphase control signal 209 for Hilbert transformation purpose, theHilbert-transformed I output 210, the Hilbert-transformed Q output 211,the complex coefficient filter I 212, the complex coefficient filter II213, the complex coefficient filter III 214, the output 215 from thecomplex coefficient filter I, and the output 216 from the complexcoefficient filter Q. Although the initial phase control signal 209 issupplied to the Hilbert transformer 207, this signal is intended tocontrol reset inputs to the flip-flops U3 and U4 shown in FIG. 11 so asto obtain I=0 and Q=0.

The configuration of elements newly added to the radio receiving systemaccording to the present invention will be described with reference toFIG. 23. Elements newly added to the circuit shown in FIG. 23 are aramping circuit 221 for gently transmitting the influence of an incomingburst-like signal at an initial stage to subsequent circuits, a firstsubtracter 222, an output 223 from the first subtracter 222, a datadelay device 224, and a second subtracter 225.

The ramping circuit 221 receives a burst-like an output from thesample-and-hold circuit 206 and supplies the thus-received signal to thefirst subtracter 222 by way of the ramping circuit 221. Upon receipt ofthe output 223 from the first subtracter 222, the delay device 224supplies the previously-sampled data set to the first subtracter 222while the sign of the data set is reversed. Accordingly, the output 223from the first subtracter 222 is gradually changed so as to indicate themean d.c. potential of the sample-and-hold circuit output 206. Theoutput 223 which indicates the mean d.c. potential is supplied to thesecond subtracter 225, where a mean d.c. potential is removed from theoutput 203 from the band pass filter 202.

From the foregoing descriptions, it is evident that the foregoingcircuit is effective in removing a d.c. component contained in a sampleoutput produced when the sample-and-hold circuit samples a receivedsignal including d.c. components or originally-unwanted components mixedin the sample output such as temperature drift of the sample-and-holdcircuit or d.c. offset of the input circuit.

(Eighth Embodiment)

Although the I signal and the Q signal are alternately input to thefirst subtracter 222 in the seventh embodiment, unnecessary calculationof a d.c. level is performed every four data sets, and hence d.c. levelsof the I and Q signals irrelevant to each other are compared with eachother with regard to two data sets prior to and subsequent to data setsof interest.

To solve the foregoing problem, a radio receiving system according to aneighth embodiment of the present invention in which a channel filter isformed by quantizing a received signal by sampling and by subjecting thethus-quantized signal to a digitized-signal processing operation,comprising: sampling means which is made of a sample-and-hold circuitand which samples the received signal; Hilbert transformation meanswhich produces rectangular components from the sample output from thesample-and-hold circuit; difference calculation means for calculating adifference between one of the rectangular components received from thetransformation means and a previously-sampled rectangular component ofthe same type; and means for calculating a difference between the outputfrom the difference calculation means and an output from the band passfilter and inputs the thus-obtained difference to the sample-and-holdcircuit.

FIG. 24 is a circuit diagram for explaining a radio receiving systemaccording to a seventh embodiment. In FIG. 23, the following elementsare the same as those shown in FIG. 2: that is, the RF signal 201, theband pass filter 202, the output 203 from the band pass filter 202, thesample-and-hold circuit 204, the sample-and-hold signal 205, thesample-and-hold circuit output 206, the Hilbert transformer circuit 207,the clock signal 298 for Hilbert transformation purpose, the initialphase control signal 209 for Hilbert transformation purpose, theHilbert-transformed I output 210, the Hilbert-transformed Q output 211,the complex coefficient filter I 212, the complex coefficient filter II213, the complex coefficient filter III 214, the output 215 from thecomplex coefficient filter I, and the output 216 from the complexcoefficient filter Q. Although the initial phase control signal 209 issupplied to the Hilbert transformer 207, this signal is intended tocontrol reset inputs to the flip-flops U 3 and U 4 shown in FIG. 11 soas to obtain I=0 and Q=0.

The configuration of elements newly added to the radio receiving systemaccording to the present invention will be described with reference toFIG. 23. Elements newly added to the circuit shown in FIG. 23 are aramping circuit 221 for gently transmitting the influence of an incomingburst-like signal at an initial stage to subsequent circuits, a firstsubtracter 222, an output 223 from the first subtracter 222, a datadelay device 224, and a second subtracter 225.

The ramping circuit 221 receives a burst-like an output from thesample-and-hold circuit 206 and supplies the thus-received signal to thefirst subtracter 222 by way of the ramping circuit 221. Upon receipt ofthe output 223 from the first subtracter 222, the delay device 224supplies the previously-sampled data set to the first subtracter 222while the sign of the data set is reversed. Accordingly, the output 223from the first subtracter 222 is gradually changed so as to indicate themean d.c. potential of the sample-and-hold circuit output 206. Theoutput 223 which indicates the mean d.c. potential is supplied to thesecond subtracter 225, where a mean d.c. potential is removed from theoutput 203 from the band pass filter 202.

From the foregoing descriptions, it is evident that the foregoingcircuit is effective in removing a d.c. component contained in a sampleoutput produced when the sample-and-hold circuit samples a receivedsignal including d.c. components or originally-unwanted components mixedin the sample output such as temperature drift of the sample-and-holdcircuit or d.c. offset of the input circuit.

(Eighth Embodiment)

The I signal and the Q signal are alternately input to the firstsubtracter 222 in the seventh embodiment. More specifically, the d.c.level of the input signal must be calculated every four data sets. Thed.c. level of a data set before the input signal and that of the inputsignal and the d.c. level of a data set subsequent to the input signaland that of the input signal are compared with each other.

To solve the foregoing problem, an eighth embodiment is directed to aradio receiving system in which a channel filter is formed by quantizinga received signal through sampling and by subjecting the thus-quantizedsignal to a digitized-signal processing operation, further comprising:

sampling means which is made of a sample-and-hold circuit and whichsamples the received signal;

Hilbert transformation means which produces rectangular components fromthe sample output from the sample-and-hold circuit;

difference calculation means for calculating a difference between one ofthe rectangular components received from the transformation means and apreviously-sampled rectangular component of the same type; and

means for calculating a difference between the output from thedifference calculation means and an output from the band pass filter andinputs the thus-obtained difference to the sample-and-hold circuit.

FIG. 24 is a circuit diagram for explaining a radio receiving systemaccording to an eighth embodiment. In FIG. 24, the following elementsare the same as those shown in FIG. 2 or 23: that is, the RF signal 201,the band pass filter 202, the output 203 from the band pass filter 202,the sample-and-hold circuit 204, the sample-and-hold signal 205, thesample-and-hold circuit output 206, the Hilbert transformer circuit 207,the clock signal 298 for Hilbert transformation purpose, the initialphase control signal 209 for Hilbert transformation purpose, theHilbert-transformed I output 210, the Hilbert-transformed Q output 211,the complex coefficient filter I 212, the complex coefficient filter II213, the complex coefficient filter III 214, the output 215 from thecomplex coefficient filter I, and the output 216 from the complexcoefficient filter Q.

The configuration of elements newly added to the radio receiving systemaccording to the present invention will be described with reference toFIG. 24. Elements newly added to the circuit shown in FIG. 24 are aramping circuit 231 for gently transmitting the influence of an incomingburst-like signal at an initial stage to subsequent circuits, a firstsubtracter 232, an output 233 from the first subtracter 232, a datadelay device 234, and a second subtracter 235.

The Hilbert-transformed I output 210 or the Hilbert-transformed Q output211 is supplied to the first subtracter 232 via the ramping circuit 231.Upon receipt of the output 233 from the first subtracter 232, the delaydevice 234 supplies the previously-sampled data set to the firstsubtracter 232 while the sign of the data set is reversed. Accordingly,the output 233 from the first subtracter 232 is gradually changed so asto indicate the mean d.c. potential of the sample-and-hold circuitoutput 206. The output 233 which indicates the mean d.c. potential issupplied to the second subtracter 235, where a mean d.c. potential isremoved from the output 203 from the band pass filter 202. Since thesignal is specified to either the I or Q signal, the result ofcalculation is not wasted.

From the foregoing descriptions, it is evident that the foregoingcircuit responds exactly and is effective in removing a d.c. componentcontained in a sample output produced when the sample-and-hold circuitsamples a received signal including d.c. components ororiginally-unwanted components mixed in the sample output such astemperature drift of the sample-and-hold circuit or d.c. offset of theinput circuit.

(Ninth Embodiment)

Under the direct conversion receiving method, there is a need to providea base band circuit with a function as a substitute for a channel filterwhich is conventionally provided in an IF stage of the existingreceiver. To this end, it is also necessary for an HF stage whosefiltering is insufficient to maintain a wide dynamic range and a widebandwidth. Still further, there is a need for a filter which filters asignal having such a wide dynamic range and a bandwidth.

FIGS. 7A to 7C show the theoretical characteristics of the channelfilter composed of two sets of cascaded three-stage complex coefficientfilters. The foregoing channel filter including the three-stage complexcoefficient filter causes an attenuation of −125 dB or more at thecenter frequency of the adjacent waveform. Even in the boundary regionbetween the adjacent waveforms, there is achieved an attenuation of −25dB. In other words, in view of the bandwidth of the channel frequency,four channels can be ensured between the image frequencies formed in thesampling waveform. A very efficient filter having superior symmetry canbe realized by connecting tandem the channel filters, each including theforegoing complex coefficient filter, and by diminishing the samplingfrequency to a quarter of the original sampling frequency in asubsequent stage.

FIG. 7A shows the frequency characteristics of the preliminary channelfilter including the complex coefficient filter shown in FIG. 2 when thefilter is activated at an oversampling frequency whose bandwidth is 64times as wide as a frequency bandwidth. FIG. 7B shows the frequencycharacteristics of the channel filter—which includes a subsequentthree-stage complex coefficient filter having the same structure as thatshown in FIG. 4D—when the filter is activated at a frequency whosebandwidth is 16 times as wide as that of the frequency bandwidth. FIG.7C shows synthetic characteristics obtained when these two channelfilters are connected tandem. It is obvious from the characteristicsshown in FIG. 7C that 16 channels are ensured between the samplingfrequencies. It is also evident that the amount of signal passingthrough the channel filter is attenuated to −30 dB or more in theboundary region between the next adjacent waveform and the adjacentwaveform after the next and that the signal is attenuated to −60 dB inthe boundary between the seventh adjacent waveform and the eighthadjacent waveform. As a matter of course, it is obvious that a nullpoint occurs in the filter at the center frequency of the adjacentwaveform and that there arises an attenuation of −25 dB or more.

From the foregoing descriptions, it is clear that the preliminarychannel filter including the three-stage complex coefficient filteractivated at a 64-times oversampling frequency has the dominant effectof attenuating the adjacent waveforms spaced frequencies away from thedesired waveform.

A ninth embodiment of the present invention has been conceived to solvethe aforementioned problem. FIG. 25 is a block diagram for explainingthe ninth embodiment. In an integrated circuit including elements whichare manufactured with limited accuracy, the precision of a preliminarychannel filter activated in a prioritized manner through high-ordersampling is improved. As a result, the effect of attenuating an adjacentwaveform spaced frequencies away from the desired waveform is moreefficiently improved. A channel filter circuit provided in a subsequentstage can receive a signal of desired waveform from only the vicinity ofwhich adjacent waveform signals of strong level are removed, i.e., asignal whose dynamic range is reduced to a lower level. Consequently, ifa channel filter provided on a subsequent stage is inferior in precisionto a channel filter provided on a preceding stage, or if the subsequentchannel filter has the insufficient capability of eliminating anadjacent waveform of great level, the radio receiving system becomesless apt to suffer from a problem.

(Tenth Embodiment)

In the case of the integrated circuit which includes elementsmanufactured with limited accuracy and which is described for the ninthembodiment, it is not easy to improve the precision of a preliminarychannel filter activated in a prioritized manner through high-ordersampling. The reason for this is that, in a case where a received signalis received in the form an analog discrete signal and the thus-receivedsignal is calculated through use of an analog circuit, means whichdetermines the accuracy of calculation is determined by the precision ofmanufacture of a circuit device. It is impossible for all the circuitdevices to achieve accuracy equivalent to −60 dB or a high accuracy ofgreater than 1/1000. For example, in the case of an analog circuit, witha view to realizing high accuracy through use of a switched capacitor,it is required to manufacture the capacitor with high accuracy of 1/1000or greater. In the case of a common capacitor, the greater the areas ofopposing metal plates become greater, the higher the accuracy ofmanufacture of the capacitor is increased. In short, when a capacitorhaving smaller electrostatic capacitance is manufactured, the accuracyof manufacture of the capacitor is decreased.

Since the complex coefficient filters do not have the same coefficient,as a matter of course, there is a need for a capacitor having smallelectrostatic capacitance. In other words, the accuracy of calculationis dominantly determined by an error in the capacitance of the capacitorhaving the minimum electrostatic capacitance.

Since a preliminary channel filter activated at a high-orderoversampling frequency responds to a sampling operation higher than thatto which a subsequent channel filter responds, there is a need for thepreliminary filter having smaller capacitance than that of thesubsequent filter. In short, the capacitor of the preliminary channelfilter is required to have capacitance smaller than that of thecapacitor of the subsequent channel filter.

A tenth embodiment of the present invention has been conceived to solvethe foregoing problem. More specifically, with a view to realizing acapacitor having small capacitance and improved accuracy, the capacitoris realized by connecting in series capacitors having capacitancegreater than that of the desired capacitor. In short, the dimensionalaccuracy of the capacitor is improved through use of a capacitor whichcan be manufactured with high accuracy, and a capacitor having smallercapacitance is realized by connecting in series a plurality ofcapacitors whose dimensional accuracy can be ensured. FIGS. 26A and 26Bshow their examples. These drawings are intended to explain a radioreceiving system according to a tenth embodiment of the presentinvention. FIG. 26A is a schematic representation for explaining theprinciple on which an error arises during manufacture of a capacitor,and FIG. 26B is a schematic representation showing the principle of thetenth embodiment.

FIG. 26A-1 shows the shape of an electrode used in manufacturing acapacitor having capacitance Co.

FIG. 26A-2 shows the shape of an electrode used in manufacturing acapacitor having capacitance nCo.

Provided that n=9, that the electrodes shown in FIGS. 26A-1 and 26A-2are geometrically similar to each other, and that the dimensionalaccuracy of manufacture of a capacitor is expressed as an absolutequantity ±δ, both the electrode shown in FIG. 26A-1 and the electrodeshown in FIG. 26A-2 have an error of γδ, and a capacitance ratio of theelectrode shown in FIG. 26A-1 to the electrode shown in FIG. 26A-2 is“n.” The ratio of one side of the electrode shown in FIG. 26A-1 to thatof the electrode shown in FIG. 26A-2 corresponds to the square of “n.”In the case of n=9, the root square of 9 is three. Provided that theelectrode shown in FIG. 26A-1 has an error of 30%, the electrode shownin FIG. 26A-2 has an error of 30/3. Given that the electrode has asquare shape, the electrode shown in FIG. 26A-1 has an area error of(1±0.3) 2 and has an error of ±0.1. The electrode shown in FIG. 26A-2has an area error of (1±0.3/3) 2 and has an error of ±0.01. It isobvious that the error ratio of the electrode shown in FIG. 26A-1 to theelectrode shown in FIG. 26A-2 is 10:1.

FIG. 26B-1 shows one capacitor and FIG. 26B-2 shows n-capacitors whosetotal capacitance is the same as that of the capacitor shown in FIG.26B-1, wherein “n” is nine. Give n th at the capacitance of thecapacitor shown in FIG. 26B-1 is Co, that the capacitance of each of thecapacitors shown in FIG. 26B-2 is C1, C2, C3, C4, C5, C6, C7, C8, C9,and that C1=C2=C3=C4=C5=C6=C7=C8=C9, 9Co=C1. As can be seen from FIG.26A, although the electrostatic capacitance of the capacitor shown inFIG. 26B-1 has an error of ±10%, the error in the electrostaticcapacitance of the capacitor shown in FIG. 26B-2 is reduced to ±1%.

As mentioned previously, the frequency characteristics of the firstfilter (i.e., a band pass filter)—for the purpose of permitting passageof only a signal at a frequency band assigned to a communications systemfrom which the radio receiving system according to the first embodimentreceives a signal—are set so as to become opposite to the apertureeffect due to a sampling operation. The band pass filter having suchcharacteristics can be readily realized through use of a SAW (surfaceacoustic wave) filter.

What is claimed is:
 1. A radio receiving system, comprising: a channelfilter including an equalizer and formed by quantizing a received signalthrough sampling and by subjecting the thus-quantized signal to adigitized-signal processing operation; and a band pass filter having anoutput connected to the channel filter, said band pass filter and saidequalizer having characteristics of compensating for the aperture effectdue to a sampling operation and provided in an input receiving stage. 2.The radio receiving system as defined in claim 1, further comprising: asample-and-hold circuit for sampling and holding an output from the bandpass filter; and an integrating circuit having a function of integratingthe received signal during a period of sampling operation of thesample-and-hold circuit.
 3. The radio receiving system as defined inclaim 2, wherein integral action time of the integrating circuit can bechanged or selected from a plurality of values.
 4. The radio receivingsystem as defined in claim 3, wherein the integral capacity of theintegrating circuit is made variable.
 5. The radio receiving system asdefined in claim 2, wherein an integrating gate function of theintegrating circuit is arranged so as to produce a Nyquist signalwaveform.
 6. The radio receiving system as defined in claim 2 or 3,wherein the time constant of the sample-and-hold circuit is set so as tobecome longer than the sampling frequency.
 7. The radio receiving systemas defined in any one of claims 1 to 3, further comprising: samplingmeans which is made of a sample-and-hold circuit and which samples thereceived signal; difference calculation means for calculating adifference between a currently-sampled signal received from thesample-and-hold circuit and a previously-sampled signal; and means forcalculating a difference between the output from the differencecalculation means and an output from the band pass filter and inputs thethus-obtained difference to the sample-and-hold circuit.
 8. The radioreceiving system as defined in any one of claims 1 to 3, furthercomprising: sampling means which is made of a sample-and-hold circuitand which samples the received signal; Hilbert transformation meanswhich produces rectangular components from the sample output from thesample-and-hold circuit; difference calculation means for calculating adifference between one of the rectangular components received from thetransformation means and a previously-sampled rectangular component ofthe same type; and means for calculating a difference between the outputfrom the difference calculation means and an output from the band passfilter and inputs the thus-obtained difference to the sample-and-holdcircuit.
 9. The radio receiving system of claim 1, wherein the output ofthe band pass filter is connected to the channel filter through alow-noise HF amplifier.
 10. The radio receiving system of claim 1,wherein the equalizer compensates for phase rotation error and the bandpass filter compensates for amplitude attenuation.
 11. A radio receivingsystem, comprising: a channel filter formed by quantizing a receivedsignal through sampling and by subjecting the thus-quantized signal to adigitized-signal processing operation; and a band pass filter having anoutput connected to the channel filter, said band pass filter having aninverse filter characteristic that boosts signal amplitude within theband at frequencies different from a center frequency of the bandwhereby said band pass filter compensates for an aperture effect due toa sampling operation and said band pass filter is provided in an inputreceiving stage.
 12. The radio receiving system as defined in claim 11wherein the band pass filter has a filter characteristic that issymmetrical about the center frequency.
 13. The radio receiving systemas defined in claim 11 wherein the band pass filter has a filtercharacteristic of (πf/fs)/(sin(πf/fs)).
 14. The radio receiving systemas defined in claim 13 wherein the band pass filter is an analog filter.15. The radio receiving system as defined in claim 11 wherein theaperture effect is distortion caused by aperture characteristics of thechannel filter having a center of frequency components at zerofrequency.
 16. The radio receiving system as defined in claim 15 whereinthe aperture effect causes a loss of information after decimation.